This invention pertains generally to the field of battery charging and particularly to the high voltage charging of multiple cells or batteries connected in series.
High voltage batteries are a critical element of several important applications such as electric vehicle drives. As with any battery, charging high voltage batteries is a complex electrochemical process in which a charging system replenishes a discharged battery by supplying to it a controlled amount of energy from an electric network. Achieving wide market acceptance for high voltage battery applications demands an economically viable system for charging high voltage batteries. Addressing this demand requires developing a low cost, high power density charging system that can supply a controlled charging current at high output voltages. However, realizing such a system requires overcoming certain practical problems related to the high output voltage.
In principle, a battery charger is a power supply with controllable voltage and current limits. What differentiates a battery charger from a conventional power supply is the capability to satisfy the unique requirements of a battery. Typically, battery chargers have two tasks to accomplish. The first, and most important, is to restore capacity as quickly as possible and the second is to maintain capacity by compensating for self-discharge and ambient temperature variations. These tasks are normally accomplished by controlling the output voltage and current of the charger in a preset manner, namely, using a charging algorithm.
The two most common charging algorithms are constant-voltage charging and constant-current charging. In constant-voltage charging, the voltage across the battery terminals is held constant, with the state of the battery determining the charge current level. The charging process normally terminates after a certain time limit is reached. Constant-voltage charging is most popular in float mode applications.
By contrast, constant-current charging holds the charging current constant. This method is often used in cyclic applications as it recharges the battery in a relatively short time.
There are many variations of the two basic methods using a succession of constant-current charging and constant-voltage charging to optimize battery charge acceptance. These variations, however, require a controlled charger with both voltage and current regulation capability.
Chargers are commonly divided into uncontrolled and controlled chargers. Uncontrolled chargers are the oldest, simplest, and cheapest chargers available. They are typically less efficient and have slow dynamic response. The simplest uncontrolled charger consists of a low frequency power transformer along with an uncontrolled bridge rectifier. Such a charger is suited for constant-voltage charging, where the battery""s state of charge sets the charging current. The advantages of such chargers include simple structure and low cost. However, with these chargers, the output voltage depends on the input voltage and has considerable voltage ripple. In addition, this type of charger could cause damage to batteries because it lacks control of the charging current.
Alternatively, controlled chargers can overcome these limitations. Controlled chargers offer the ability to control the charging current as well as to implement both constant-voltage and constant-current charging methods. The simplest form of controlled chargers are SCR chargers, consisting of a low frequency transformer, an SCR bridge rectifier, and a DC choke. SCR chargers offer a simple and low cost solution to implement a fully controllable charging system. They are still in use in many low to high power industrial applications. However, SCR chargers are bulky and have relatively low efficiencies and slow dynamic response.
Transistor controlled chargers comprise another class of controlled chargers. They consist of a low frequency transformer, an uncontrolled bridge rectifier, and a series pass transistor. These chargers can implement both constant-voltage and constant-current charging methods and have fast dynamic response. However, they have low efficiencies and are generally bulky due to the low frequency transformer.
Switch mode power supply (SMPS) based chargers offer improved performance compared to the SCR and the transistor controlled chargers. These chargers offer high efficiency power conversion due to high frequency operation. The high frequency power conversion stage results in significant size reduction for the energy storage elements (transformers, inductors, and capacitors). In addition, these chargers have fast dynamic response. The basic components of an SMPS charger include an input filter stage, an input rectification stage, a power factor correction stage (if required), a high frequency power conversion power stage, a high frequency isolation transformer, and an output rectification and filtering stage. A central analog/digital controller is normally employed to regulate the charger voltage/current and to implement the desired charging algorithm. Considering that a well designed switch mode power supply is inherently current limited, the combination of constant-current and constant-voltage charge is available.
In order to implement both constant-voltage and constant-current charging methods, a SMPS charger would employ an output filtering stage that allows for output current limiting. This is typically achieved by using an inductive output filtering stage (DC choke), which smoothes the output charging current and limits it through the charger control circuitry. The inductor serves as the main energy storage device. Consequently, the charging current is smoothed out and is prevented from changing instantaneously. This allows the SMPS charger to implement accurate current limiting as well as protect against any short circuit conditions that may arise across the output terminals. The inductor current is normally sensed and regulated by the control circuitry to achieve the desired level of output current and to implement the constant-current intervals of the charging algorithm. An output capacitor is normally used to filter out any remaining current ripple in the filter inductor and thus supply a pure DC current to the battery. The voltage across the capacitor, which is the same as the battery voltage, is normally sensed and regulated by the control circuitry to achieve the desired level of output voltage and implement the constant-voltage intervals of the charging algorithm.
For low power battery charging needs ( less than 1 kW), the single switch and the two switch forward converters are the simplest isolated SMPS battery charger topologies with an inductive output filtering stage. A battery charger using a single switch forward converter power stage may employ a half wave rectifier on the secondary side.
For high power charging needs ( greater than 1 kW), the full-bridge converter of the type shown in FIG. 1 (H-bridges of transistors Q1-Q4) with an inductive output filter L0 is the most suitable power converter topology. A full wave rectifier composed of diodes D1-D4 is employed on the secondary side to rectify the primary voltage and current waveforms. Typically, a center-tapped (push-pull) or a full-bridge rectifier is used in association with a full-bridge converter topology. Typical voltage and current waveforms for the full-bridge SMPS charger are shown in FIG. 2. With an inductive output filtering stage, the secondary rectifiers are normally subjected to high voltage transients (ringing) during switching transitions. This is due to the reverse recovery of the output diodes where the transformer leakage inductance resonates with the secondary diodes"" junction capacitance causing a two per-unit voltage stress across them. With an input DC bus voltage of VDC and a transformer turns ratio of 1:a, the diode voltage stress is approximately twice the transformer secondary voltage, namely 2xc2x7axc2x7VDC. Consequently, the secondary diodes"" voltage rating should be higher than twice the secondary reflected input DC bus voltage. The reverse recovery of the secondary diodes causes additional switching losses that become more dominant at higher switching frequencies. Improving the reverse recovery behavior of the secondary diodes, then, would directly improve overall charger performance and efficiency.
In order to alleviate ringing and switching losses associated with reverse recovery of power diodes, designers have used either Schottky diodes, which have no or minimal reverse recovery, or Ultrafast diodes, which have soft reverse recovery behavior. Both of these diode technologies have been extensively employed in SMPS and charger designs to yield improved performance and higher efficiency designs. Schottky diodes, however, are commercially available only in low voltage ratings, namely below 150V. Thus, their use has been restricted to low voltage battery chargers with battery voltages of less than 36V. On the other hand, Ultrafast diodes are offered in voltages of up to 1600V, which extends their use to high voltage SMPS applications. However, higher voltage Ultrafast diodes (e.g. 1200V diodes) do typically have higher reverse recovery characteristics compared with lower voltage ones (e.g. 600V diodes), resulting in performance degradation and lower efficiency. Thus the reverse recovery characteristic can be improved by circuit topologies that enable the use of lower voltage diodes, and preferably Schottky diodes.
With the advent of electric and hybrid electric vehicles, long battery strings consisting of tens of series-connected battery modules are becoming increasingly common. For example, a typical hybrid electric bus battery may consist of forty-eight 12V-battery modules with a nominal battery output voltage of 576V. Under charge, the battery voltage can be as high as 750V. This would require a battery charger capable of supplying a charging voltage of more than 750V. Assuming a battery amp-hour capacity is 50Ahrs and a 5-hour recharge time, an 800V/10A charger would be needed. Consequently, the charger power rating is 8kW. The full-bridge SMPS charger is well suited for this power level. Although the design of an 8kW full-bridge SMPS charger may sound quite straightforward, a number of issues would need to be resolved. Due to the high voltage nature of the charger ( greater than 750V), Schottky diodes cannot be used. As an alternative, Ultrafast diodes can be considered. Since the output voltage of the charger will be higher than 750V, the minimum voltage rating of the secondary diodes (D1-D4 in FIG. 1) is 1500V, or twice the output voltage. This would dictate the use of 1600V Ultrafast diodes. However, such diodes are not very common, as they are only offered by very few manufacturers. In addition, their reverse recovery performance is not as good as the reverse recovery performance of lower voltage Ultrafast diodes. Even if the 1600V diodes were used, their switching losses would limit the maximum converter switching frequency to a relatively low frequency, resulting in a lower power density charger. It is desirable to operate at higher switching frequencies in order to reduce the size of magnetic and capacitive filter components and to obtain a higher power density charger.
One approach that reduces the required voltage rating of the secondary rectifier diodes employs voltage clamps or snubber circuits. Voltage clamps act to clamp the voltage across the secondary diodes to a level lower than twice the charger output voltage. In this approach, the excess energy stored in the transformer leakage inductance is transferred to a clamp capacitor. The clamp capacitor discharges its energy to the load through a resistor. Part of the energy stored in the clamp capacitor is transferred to the load while the rest is dissipated as heat in the resistor. Such a clamp circuit allows the use of lower voltage diodes which would normally have improved reverse recovery characteristics and hence lower losses. For the 800V/10 A charger example described earlier, the use of a clamp capacitor may allow the use of 1200V diodes instead of 1600V parts. However, even with a clamp capacitor, the energy loss due to reverse recovery and the energy loss in the clamp resistor limit the maximum operating switching frequency, which limits charger power density.
In high voltage battery chargers, the ability to use diodes having low reverse recovery characteristics significantly reduces the stresses and losses associated with reverse recovery. Since lower voltage diodes have greatly improved reverse recovery behavior (e.g. 600V hyperfast and Stealth diodes), a converter topology for a high voltage battery charger that can use such lower voltage rating diodes would certainly exhibit improved performance. One approach that allows the use of lower voltage rating diodes is to connect the diodes in series. However, such an arrangement complicates the charger design because it requires additional circuitry to ensure voltage sharing during turn off.
Therefore, what is needed is a battery charger, a method of charging, and a suitable controller that permit the use of low reverse recovery diodes in charging high voltage strings of series-connected batteries. Furthermore, what is also needed is such a high voltage battery charging system capable of simple and flexible reconfiguration for charging lower voltages at higher currents. Furthermore, such a system is needed that is controllable to provide both constant-current charging and constant-voltage charging.
In accordance with the present invention, a high voltage battery charger is provided for efficiently and economically charging high voltage strings of series-connected batteries. The present invention provides for a plurality of secondary output terminals that can be arranged in series or parallel networks to adapt to a variety of output voltage and current levels. Reconfiguring the charger from one network arrangement to another requires simply adjusting a voltage feedback scale factor and rearranging the connections between the secondary output terminals and the battery charger output terminals. The battery charger of the invention can be controlled to operate selectively in either constant voltage or constant current mode charging.
The battery charger in accordance with the present invention includes a buck-based DC-to-AC converter circuit comprising an arrangement of controllable switching devices, having its input connected to a DC source and its output connected to one or more transformer primaries. The transformer may have a single primary and a plurality of secondary windings having preferably equal numbers of turns. Each secondary winding is connected to a corresponding secondary circuit. Each secondary circuit includes a rectification circuit connected across a respective secondary winding, where the AC input terminals of each rectification circuit are connected to the terminals of each corresponding secondary winding. The rectification circuit has a pair of DC output terminals connected to a low-pass L-C output filter, comprising an output inductor and an output capacitor. The positive and negative terminals of each output capacitor can be connected in either series or parallel networks to provide a range of output capabilities, including both maximum voltage capability and maximum current capability. Further, the battery charger includes a positive output terminal and a negative output terminal to provide for connection to a string of batteries to be charged. Within the network of output capacitors, the most positive output capacitor terminal connects to the positive output terminal and the most negative output capacitor terminal connects to the negative output terminal.
The controllable switching devices in the DC-to-AC converter can be any suitable devices known to those skilled in the art, but are preferably power MOSFETs or IGBTs. The DC-to-AC converter circuit is preferably a full-bridge inverter, but may alternatively comprise a half-bridge inverter or a forward converter. In a preferred embodiment, one branch of the DC-to-AC converter connects to a terminal of the primary winding through a DC blocking capacitor to prevent saturation of the transformer due to asymmetric inverter operation. The DC source can be any suitable DC voltage source, but preferably comprises a DC capacitor fed by a rectifier with its inputs connected to an AC source, such as, for example, the utility mains. The transformer can include any feasible number of secondary windings, but a transformer designed for charging a typical electric vehicle battery preferably has about six secondary windings. The rectification circuit in the preferred embodiment is preferably a full-wave bridge rectifier, but may be half-wave or free-wheeling diode networks adapted to operate in conjunction with the applied DC-to-AC converter circuit. The rectification circuit comprises a network of any suitable diodes, and preferably diodes having low reverse recovery characteristics, such as Schottky diodes, Ultrafast diodes, or Stealth diodes, for example. The output capacitors may be all parallel-connected, all series-connected, or connected in a plurality of strings of series-connected output capacitors wherein the strings of series-connected output capacitors are connected in parallel.
A battery charger controller in accordance with the present invention provides signals to control the controllable switching devices in the DC-to-AC converter. The controller provides an output signal on an output line to a pulse generator and gate drive circuit to cause the controllable switching devices in the DC-to-AC converter to switch on and off periodically at a selected frequency to couple energy from the primary winding to the circuits connected to the secondary windings. The controller utilizes a set of feedback signals proportional to the average current in the output inductors. In one embodiment, a summing circuit in the controller adds the signals in the set of feedback signals together to form an average current feedback signal representing the average of the currents flowing in the individual output inductors. The controller also accepts a signal comprising the voltage across the output terminals, which is preferably scaled to control signal levels by an adjustable voltage feedback scaling factor, kv.
The controller regulates the scaled signal representing the voltage across the output terminals to a commanded value, Vo_limit. To protect against over-current conditions, the controller compares the average current feedback signal to a specified current limit that establishes the maximum output current desired by the user. When the average current feedback signal is below the specified upper current limit the controller operates in constant-voltage mode and the controller operates in constant-current mode when the average current feedback signal exceeds the specified upper current limit.
The controller regulates the average current feedback signal using an inner control loop having a first control bandwidth, and regulates the voltage across the output terminals using an outer control loop at a second control bandwidth that is less than the first control bandwidth. Preferably, the second control bandwidth is less than or equal to 10% of the first bandwidth. The inner control loop of the controller further preferably includes a proportional-integral (PI) compensation network. The outer control loop of the controller also preferably includes a PI compensation network.
The present invention includes a current sensor to monitor each output inductor current signal in the set of feedback signals; these current signals are then summed in a summing circuit in the controller. The set of feedback signals to the controller can comprise signals formed by current sensors that accept windings of at least one turn from conductors in series with each output inductor, such as Hall effect current sensors. Preferably, the set of feedback signals is replaced by a single output signal from a current sensor comprising a core having at least one turn of each output inductor current. In the preferred embodiment, only a single turn of each output inductor current is used, and the resulting current sensor output signal represents the scaled sum of output inductor currents, the scaling depending on the number of turns of the sense winding. In this manner, the need for a summation circuit can be avoided.
Further objects, features, and advantages of the present invention will be apparent from the following detailed description when taken in conjunction with the accompanying drawings.